Methods and apparatus for wirelessly communicating signals that include embedded synchronization/pilot sequences

ABSTRACT

An embodiment for wirelessly communicating a signal includes a transmitter combining a plurality of phase shifted input data signals with a plurality of synchronization/pilot sequences to produce a plurality of combined signals, performing frequency domain-to-time domain transformations of the combined signals to produce a plurality of candidate signals, determining peak-to-average ratios for at least some of the plurality of candidate signals, identifying a selected signal from the plurality of candidate signals based on the peak-to-average ratios, and transmitting the selected signal over a wireless communication channel. An embodiment further includes a receiver receiving a channel-affected version of the transmitted, selected signal, determining an estimate of a selective mapping index, which identifies the synchronization/pilot sequence from the plurality of synchronization/pilot sequences, applying corrections to the received signal based on estimated channel perturbations to produce an equalized combined signal, and producing an output data symbol from the equalized combined signal.

CROSS-REFERENCE TO RELATED APPLICATION

This application claims the benefit of U.S. Provisional Application No.60/911,787, filed Apr. 13, 2007.

GOVERNMENT LICENSE RIGHTS

The U.S. Government may have certain rights to some or all of theinventive subject matter of the present application as provided for bythe terms of contract No. DAAD19-01-2-0011 awarded by Army ResearchLaboratory.

TECHNICAL FIELD

The inventive subject matter generally relates to methods and apparatusfor wirelessly communicating signals, and more particularly to methodsand apparatus for wirelessly communicating signals that include embeddedsynchronization/pilot sequences in multi-carrier communication systems.

BACKGROUND

In multi-carrier communication systems, orthogonal frequency divisionmultiplexing (OFDM) is an effective, high-speed communications techniquethat allows for relatively efficient multi-path channel equalization.However, signals generated using traditional OFDM techniques tend tosuffer from relatively large peak-to-average ratios (PARs) orpeak-to-average power ratios (PAPRs), which in turn may lead tosignificant distortion noise and low power efficiency in peak-limitedchannels. In addition, under relatively harsh channel conditions,transmitted OFDM signals tend to incur significant timing offsets andcarrier frequency offsets. Because traditional OFDM techniques tend notto be robust under harsh channel conditions, significant timing offsetsmay result in inter-block interference, and significant carrierfrequency offsets may result in inter-carrier interference. Both ofthese forms of interference are detrimental to the bit error rates ofreceived signals.

In order to mitigate these detrimental effects, some traditional OFDMmethods include, on the transmitter side, transmitting a synchronizationand/or channel estimation preamble in conjunction with and precedingeach transmit information sequence. On the receiver side, the preambleis used during signal acquisition to synchronize to the received signaland, when the preamble also includes channel training information, italso may be used to perform channel estimation (e.g., estimatingtransmission channel parameters such as timing offset, carrier frequencyoffset, and multi-path fading). Although transmission of a preamble isrelatively simple to implement, a tradeoff to implementing thistechnique is that a significant amount of bandwidth is used solely forpreamble transmission, and thus for synchronization, acquisition, and,when channel training information is available, also for channelestimation. Furthermore, using preamble-based techniques, there are nooptions for re-acquiring a signal during the information sequencereception portion of the transmission if synchronization is lost afterthe preamble is received.

Another traditional OFDM method excludes the use of a preamble forsynchronization, and instead a cyclic prefix (or cyclic extension) isincluded within each transmitted OFDM symbol. Using a cyclic prefix, thefirst G samples represented in the OFDM symbol are an exact copy of thelast G samples of the inverse fast Fourier transform (IFFT) basebandoutput. The cyclic prefix can be used for timing and frequencysynchronization, but this method is generally less robust than usingmethods that include transmission of a preamble because G typically isselected to be less than the length of a typical preamble (e.g., apreamble having length L). Comparatively, for example, if L≈10G, thedetection performance (e.g., the detection signal-to-noise ratio (SNR))should be about 10 dB higher using a preamble of length L compared to acyclic prefix of length G. Because each OFDM symbol contains a cyclicprefix, a correlation may be performed on each OFDM symbol, and then thecorrelations may be integrated over multiple symbols (e.g., 10 OFDMsymbols) in order to achieve substantially equivalent detection SNR.However, this incurs a processing delay proportional to the number ofOFDM symbols over which the correlations are integrated (e.g., aprocessing delay of 10 OFDM symbols), which is unacceptable in manyapplications. In addition, a cyclic prefix is not useful for PARreduction.

In order to mitigate the effects of the channel on a received signal,some traditional OFDM methods also perform channel estimation, on thereceiver side. The channel estimate may be used to compensate forclipping associated with a limited channel dynamic range, timingoffsets, carrier frequency offsets, and multi-path fading among otherthings. Naturally, the channel estimate has some error, when comparedwith actual channel conditions. Traditional OFDM transmission methodsmay experience an increase in channel estimation errors on the receiverside, which may result from non-linear amplification, by a poweramplifier device on the transmitter side, of transmit informationsequences having higher than desired peak-to-average power ratios. Suchnon-linear transmission may cause significant out-of-band interference(i.e., interference outside the signal bandwidth, such as in theadjacent channels and/or other user channels), and also may induceundesired in-band interference, which adds distortion the transmittedinformation bits and also to the channel training information.Furthermore, improper synthesis of the channel training information maylead to further channel estimation errors at the receiver. Thus,non-linear amplification of high peak-to-average power ratio signals andimproper channel training information design may, in the receiver,result in unacceptably high channel estimation errors and excessivelyhigh bit error rates.

Accordingly, what are needed are methods and apparatus for communicatingwireless signals in multi-path communication systems, in whichsynchronization and acquisition are performed in a bandwidth-efficientmanner that is relatively robust under harsh channel conditions. Alsoneeded are methods and apparatus that enable re-acquisition of a signalto be performed if synchronization is lost during the informationsequence reception portion of a transmission. Also needed are methodsand apparatus for communicating wireless signals having improvedpeak-to-average power ratios and/or bit error rates, when compared withtraditional methods and apparatus. Other features and characteristics ofthe inventive subject matter will become apparent from the subsequentdetailed description and the appended claims, taken in conjunction withthe accompanying drawings and this background.

BRIEF DESCRIPTION OF THE DRAWINGS

The inventive subject matter will hereinafter be described inconjunction with the following drawing figures, wherein like numeralsdenote like elements, and

FIG. 1 is a simplified block diagram of a multi-carrier communicationsystem that includes multiple wireless communication devices thatcommunicate over a wireless communication channel, in accordance with anexample embodiment;

FIG. 2 is a simplified block diagram of a channel model, in accordancewith an example embodiment;

FIG. 3 is a simplified block diagram of a transmitter, in accordancewith an example embodiment;

FIG. 4 is an example of a frequency-domain representation of atransmitted signal, in accordance with an example embodiment;

FIG. 5 is a simplified block diagram of a receiver, in accordance withan example embodiment;

FIG. 6 is a flowchart of a method for generating and transmittingwireless signals that include embedded synchronization/pilot sequences,in accordance with an example embodiment;

FIG. 7 is a flowchart of a method for receiving and processing wirelesssignals that include embedded synchronization/pilot sequences, inaccordance with an example embodiment;

FIG. 8 is a chart plotting bit error rate (BER) performance that may beachieved using two example synchronization/pilot sequence, selectivemapping (SPS-SLM) embodiments; and

FIG. 9 is a chart comparing peak-to-average ratio (PAR) characteristicsfor signals produced using various embedding factors and candidatenumber quantities, in accordance with various example embodiments.

DETAILED DESCRIPTION

The following detailed description of the inventive subject matter ismerely exemplary in nature and is not intended to limit the inventivesubject matter or the application and uses of the inventive subjectmatter. Furthermore, there is no intention to be bound by any theorypresented in the following detailed description.

Embodiments include methods and apparatus for wirelessly communicatingorthogonal frequency division multiplexing (OFDM) signals betweenwireless communication devices. Signals communicated according tovarious embodiments include embedded synchronization/pilot sequences, aswill be described in detail below. The embodiments may have one or moresignificant advantages over traditional OFDM techniques, in that theembodiments address the issues of signal distortion relating tonon-linear amplification of transmit information sequences having higherthan desired peak-to-average ratios (PARs), carrier frequency offsetsensitivity, and timing synchronization, among other things.

Embodiments may include PAR reduction and embedded synchronizationmethods and apparatus, and may be referred to herein as synchronizationpilot sequence (SPS) selected mapping (SLM), or SPS-SLM. SPS-SLMembodiments described herein simultaneously consider a plurality ofchannel impairments (e.g., channel perturbations) in an OFDM system, andapply a unified approach for correcting the impairments. Bysimultaneously addressing multiple channel impairments, according tovarious embodiments, overhead devoted to channel impairment correctionand/or PAR reduction may be reduced when compared with traditionalmethods that address these issues individually. This may lead to greaterbandwidth efficiencies. In addition, the various embodiments may becapable of performing better, in harsh channel environments, thantraditional OFDM techniques.

FIG. 1 is a simplified block diagram of a multi-carrier communicationsystem 100 that includes multiple wireless communication devices 102,104 that communicate over a wireless communication channel 106, inaccordance with an example embodiment. Wireless communication devices102, 104 may include, for example but not by way of limitation, a deviceselected from a group of devices comprising a cellular telephone, aradio, a one-way or two-way pager, a personal data assistant, a computer(e.g., a laptop or desktop computer), a base station, an unmannedautonomous vehicle, a wireless transmitter, and/or a wirelesstransceiver.

Embodiments may be implemented in wireless communication devices 102,104 that include both a transmitter 110, 112 and a receiver 114, 116(e.g., each device 102, 104 includes a transceiver). In suchembodiments, system 100 may provide for two-way communications betweendevices 102, 104. For example, transmitter 110 in a first device 102 mayreceive an input data symbol 118, X[n], and may generate and transmit,over channel 106, a wireless signal 120, y[n], which represents theinput data symbol 118. Receiver 114 in a second device 104 may receive achannel-affected version 122, z[n], of the wireless signal 120, and maygenerate an output data symbol 124, {circumflex over (X)}[n],representing an estimate of the input data symbol 118. Additionally,transmitter 112 in the second device 104 may receive another input datasymbol 130, and may generate and transmit, over channel 106, a wirelesssignal 132 representing the input data symbol. Receiver 116 in the firstdevice 102 may receive a channel-affected version 134 of the wirelesssignal 132, and may generate an output data symbol 136 representing anestimate of the input data symbol 130. In other embodiments, system 100may provide for one-way communications. For example, one device mayinclude a transmitter (and no receiver) and another device may include areceiver (and no transmitter). In order to more clearly and simplydescribe the various embodiments, only one-way communications between atransmitter 110 in a first device 102 and a receiver 114 in a seconddevice 104 is described in detail in the remainder of this description.It is to be understood that the various embodiments also apply totwo-way communications as well.

Functionality of transmitter 110 and receiver 114, are described onlybriefly in conjunction with the description of FIG. 1. More detaileddescriptions of the details of various transmitter and receiverembodiments are included later, in conjunction with FIGS. 3-6. Briefly,transmitter 110 is adapted to apply multiple phase shifts to an inputdata symbol 118, and to combine a plurality of synchronization/pilotsequences (SPSs) with the phase shifted input data in order to produce aplurality of candidate signals. First and second scaling factors may beapplied to the input data symbol and to the plurality of SPSs,respectively, prior to combining the phase shifted input data and theplurality of SPSs. As will be discussed in detail later, the scalingfactors affect the relative signal power allocated to the phase shiftedinput data and the SPSs with which they are combined. Transmitter 110also is adapted to determine PARs for at least some of the candidatesignals, and to identify a selected candidate signal based on the PARs(e.g., the selected candidate signal may be the candidate signal withthe lowest PAR). Transmitter 110 also is adapted to transmit a wirelesssignal 120 representing the selected candidate signal over the wirelesscommunication channel 106.

Receiver 114 is adapted to receive a channel-affected version 122 of thewireless signal 120 from the wireless communication channel 106.Receiver 114 also is adapted to determine estimated channelperturbations within the channel-affected signal 122 based on itsknowledge of the plurality of SPSs, and to apply corrections to thechannel-affected signal 122, based on the estimated channelperturbations. Receiver 114 also is adapted to produce the output datasymbol 132 based on the corrected signal, which represents an estimateof the input data symbol 130 processed at the transmitter 110.

As alluded to above, a wireless signal transmitted over a channel (e.g.,channel 106) may be adversely affected by the channel, and a receiverthat receives a channel-affected version of the transmitted signal mayattempt to determine and correct for estimated channel perturbationsreflected within the channel-affected signal. In fact, the channelperturbations generated by channel 106 may not be the same for signalsfrom transmitter 110 to receiver 114 as compared to a transmission fromtransmitter 112 to receiver 116. A number of factors may inducedifferences in the forward and reverse directions. For example, wheneither or both devices 102, 104 are mobile, channel 106 will be timevariant, and the time that transmitter 110 transmits to receiver 114 maybe different from the time than transmitter 112 may transmit to receiver116. Thus, the channel 106 will be different depending on the transmittime for each transmitter 110, 112. Furthermore, the channel 106 itselfmay have different characteristics in the forward direction as comparedto the reverse direction. These differences may be induced by a numberof factors which include, for example, device 102 possessing atransmit/receive antenna having different characteristics from thetransmit/receive antenna of device 104, and/or the local scatteringenvironment being different for each device 102, 104, among otherthings. In order better to convey how a receiver may determine andcorrect for estimated channel perturbations, in accordance with variousembodiments, a simplified channel model will now be described.

FIG. 2 is a simplified block diagram of a channel model 200, inaccordance with an example embodiment. In particular, channel model 200illustrates various channel characteristics that may affect (e.g.,perturb) a signal transmitted over the channel, and more particularly anunsynchronized mobile channel that communicates signals generated by apeak power-constrained system. These characteristics include, forexample, a multi-path fading component 202 (which, in the frequencydomain, manifests itself as frequency selective fading), a timing offset(TO) component 204, a carrier frequency offset (CFO) component 206, andan additive noise component 208. Although not strictly part of thechannel model, the input-to-output characteristic of the transmitter'spower amplifier (e.g., power amplifier 316, FIG. 3), which may or maynot be assumed to be time-invariant, also may affect the characteristicsof a transmitted wireless signal. A signal, z[n], to which the channelmodel 200 and the power amplifier input-to-output characteristic hasbeen applied may be represented, for example, by the equation:

z[n]=(f _(PA)(y[n−n ₀])*h[τ])e ^(−j2πε/N) +η[n],  (Equation 1)

where f_(PA)(.) represents the power amplifier input-to-outputcharacteristic, which may be assumed to be time-invariant, h[τ]represents multi-path fading component 202, y[n−n₀] represents atransmitted signal, y[n], subjected to a TO component 204, e^(−j2πε/N)represents a CFO component 206, η[n] represents an additive noisecomponent 208, and * is the convolution operator.

More detailed descriptions of various embodiments of transmitters (e.g.,transmitter 110, FIG. 1) and receivers (e.g., receiver 114, FIG. 1) andmethods of their operation will now be described. In particular, FIG. 3is a simplified block diagram of a transmitter 300, in accordance withan example embodiment. Transmitter 300 includes a data/scaling factorcombiner 302, a plurality of phase shifters 304, a plurality ofSPS/scaling factor combiners 306, a plurality of data/SPS combiners 308,a plurality of frequency domain-to-time domain (FD-to-TD) transformers310, a signal selector 312, an up-converter 314, a power amplifier 316,and an antenna 318 operatively coupled together as illustrated in FIG.3, in an embodiment.

Data/scaling factor combiner 302 includes computational apparatusadapted to receive a sequence of input data symbols 320, X_(k), each ofwhich represents a data-bearing part of a signal to be transmitted. Inan embodiment, X_(k) is drawn from a finite constellation. Data/scalingfactor combiner 302 is further adapted to apply a first scaling factor322 to an input data symbol 320 in order to produce a scaled input datasymbol 324. In an embodiment, the first scaling factor 322 has a valueof √{square root over (1−ρ)}, where ρ is an embedding factor having avalue between 0 and 1. In a particular embodiment, the embedding factorhas a value in a range of about 0.25 to about 0.35. In anotherembodiment, the embedding factor has a value in a range of about 0.2 toabout 0.4. In still other embodiments, the embedding factor may havehigher or lower values than the above-given ranges. The scaled inputdata symbol 342 may be represented as √{square root over (1−ρ)}X_(k).

Each of the plurality of phase shifters 304 includes computationalapparatus adapted to apply a different phase shift 326, e^(jφ) ^(k)^((d)) to the scaled input data symbol 324, in order to produce aplurality of phase shifted input data signals 328, √{square root over(1−ρ)}X_(k)e^(jφ) ^(k) ^((d)) , where D is a value referred to herein asa candidate number quantity, d is an index referred to herein as arelational index, and dε{1, 2, . . . , D}. The candidate numberquantity, D, may be selected as any integer number from 1 to 16, in anembodiment, although the candidate number quantity may be a largernumber, in other embodiments. In a particular embodiment, the candidatenumber quantity is selected as an integer number between 3 and 10. In anembodiment, the number of phase shifted input data signals 328 producedequals the candidate number quantity D, although the number of phaseshifted input data signals 328 may be different, in other embodiments.The different phase shifts 326 may be represented within entries of atable of phase shift values, in an embodiment, and the relational index,d, may be used an index into the phase shift value table, among otherthings. Accordingly, the phase shift value table may have D entries, inan embodiment, although the phase shift value table may have more orfewer entries in other embodiments.

Transmitter 300 also is adapted to obtain a plurality of SPS 332, S_(k)^((d)), each of which represents a unique synchronization/pilotsequence. In an embodiment, the plurality of SPS 332 may be obtainedfrom a table of SPSs, which is accessible to or stored in transmitter300, and which includes a set of pre-generated SPSs, each of which maybe referenced by a unique index (referred to below as an SLM index).Each SPS 332 in the transmitter's SPS table is represented in thefrequency domain, in an embodiment.

SPS/scaling factor combiners 306 include computational apparatus adaptedto apply second scaling factors 330 to the plurality of SPS 332 in orderto produce a plurality of scaled SPS 334, √{square root over (ρ)}S_(k)^((d)), where d is the relational index. Similar to its functionalitywith respect to the phase shift value table, the relational index, d,also may be used an index into the SPS table. When used in this context,the relational index alternatively may be referred to as an SLM index.As with the phase shift value table, the SPS table also may have Dentries, although the SPS table may have more or fewer entries in otherembodiments. In addition, in an embodiment, the number of scaled SPS 334produced equals the candidate number quantity D, although the number ofSPS 334 may be different, in other embodiments.

In the above-described embodiment, each different phase shift value 326may be related to a unique SPS 332 via the relational index, d. Inalternate embodiments, a particular phase shift value 326 may be relatedto multiple unique SPS 332, or a particular unique SPS 332 may berelated to multiple phase shift values 326 (e.g., by including duplicatevalues in the phase shift value table or the SPS table, for example).

In an embodiment, the second scaling factor 330 has a value of √{squareroot over (ρ)}, where ρ is the same embedding factor as the embeddingfactor incorporated in the first scaling factor 322. As will be moreclearly depicted in conjunction with FIG. 4, later, because the firstand second scaling factors 322, 330 have an inverse relationship, thevalue of the embedding factor, ρ, dictates how much relative signalpower is allocated to a data-bearing component, X_(k) ^((d)), of atransmitted signal as opposed to an SPS component, S_(k) ^((d)), of thetransmitted signal.

Phase shifters 304 provide the plurality of phase shifted input datasignals 328 to data/SPS combiners 308, and SPS/scaling factor combiners306 provide the plurality of scaled SPS 334 to data/SPS combiners 308.Each of data/SPS combiners 308 includes computational apparatus adaptedto combine one of the plurality of phase shifted input data signals 328with one of the scaled SPS 334 in order to produce a plurality ofcombined signals 340, where the plurality of combined signals 340 may berepresented in the frequency domain by the equation:

Y _(k) ^((d))=√{square root over (ρ)}S _(k) ^((d))+√{square root over(1−ρ)}X _(k) ^((d)) e ^(jφ) ^(k) ^((d)) .  (Equation 2)

In an embodiment, the number of combined signals 340 produced equals thecandidate number quantity D, although the number of combined signals 340may be different, in other embodiments.

Data/SPS combiners 308 provide the plurality of combined signals 340 toFD-to-TD transformers 310. FD-to-TD transformers 310 includecomputational apparatus adapted to perform frequency domain-to-timedomain conversions on each of the combined signals 340, in order toproduce a plurality of candidate signals 342, y^((d))[n]. In anembodiment, the number of candidate signals 342 produced equals thecandidate number quantity D, although the number of candidate signals342 may be different, in other embodiments. The frequency domain-to-timedomain conversion may include an inverse Fourier transform (IFT) or,more particularly, an inverse discrete Fourier transform (IDFT), invarious embodiments, although other types of frequency domain-to-timedomain conversions may be performed in other embodiments. Accordingly,in an embodiment, the plurality of candidate signals 342 may berepresented as y^((d))[n]=IDFT{Y_(k) ^((d))} or alternatively by thefollowing:

$\begin{matrix}\begin{matrix}{{y^{(d)}\lbrack n\rbrack} = {\frac{1}{\sqrt{N}}{\sum\limits_{k = 0}^{N - 1}{Y_{k}^{(d)}^{j\; 2\; \pi \; {{kn}/N}}}}}} \\{= {{{x^{(d)}\lbrack n\rbrack}\sqrt{\left( {1 - \rho} \right)}} + {{s^{(d)}\lbrack n\rbrack}\sqrt{(\rho)}}}}\end{matrix} & \left( {{Equation}\mspace{14mu} 3} \right)\end{matrix}$

where x^((d))[n]=IDFT{X_(k)e^(jφ) ^(k) ^((d)) }, s^((d))[n]=IDFT{S_(k)^((d))}, and nε{0, 1, . . . , N−1}. In an embodiment, an efficientalgorithm for computing the inverse discrete Fourier transform (IDFT)may be implemented, such as an inverse fast Fourier transform (IFFT),for example.

The above description indicates that, in an embodiment, transmitter 300includes a number of phase shifters 304, a number of SPS/scaling factorcombiners 330, a number of data/SPS combiners 308, and a number ofFD-to-TD transformers 310 that is equal to the candidate numberquantity, D, and that these transmitter elements are adapted to generatea same number, D, of phase shifted input data signals 328, scaled SPSs334, combined signals 340, and candidate signals 342, respectively. Inother embodiments, transmitter 300 may include more or fewer than thecandidate number quantity, D, of phase shifters 304, SPS/scaling factorcombiners 330, data/SPS combiners 308, and/or FD-to-TD transformers 310,and/or some or all of these transmitter elements may be adapted togenerate more or fewer than the candidate number quantity, D, of phaseshifted input data signals 328, scaled SPSs 334, combined signals 340,and/or candidate signals 342, respectively. Although the number of phaseshifters 304, SPS/scaling factor combiners 330, data/SPS combiners 308,and/or FD-to-TD transformers 310 may be the same, in an embodiment, inother embodiments, the numbers of these transmitter components 304, 330,308, 310 and/or signals 328, 334, 340, 342 may be different. Forexample, but not by way of limitation, data/SPS combiners 308 maycombine a same phase shifted input data signal 328 with multiple scaledSPSs 334 or data/SPS combiners 308 may combine a same scaled SPS 334with multiple phase shifted input data signals 328, in variousembodiments. In other embodiments, some signals may be disregarded when,for example, they fail to meet certain criteria and/or threshold levels,which ultimately may result in fewer than the candidate number quantity,D, of candidate signals 342 being provided to signal selector 312.Accordingly, embodiments of the inventive subject matter are not limitedto there being a same number, D, of transmitter components 304, 330,308, 310 and/or signals 328, 334, 340, 342.

FD-to-TD transformers 310 provide the plurality of candidate signals 342to signal selector 312. In an embodiment, signal selector 312 includescomputational apparatus adapted to determine peak-to-average ratios(PARs) for some or all of the candidate signals 342, and based on thePARs, to identify a selected signal 346 from the candidate signals 342.

As used herein, the term peak-to-average ratio (PAR) means a measurementof a waveform that equals the peak amplitude of the waveform divided bythe root mean squared (RMS) or time averaged value of the waveform.Although PAR reduction is discussed extensively herein, embodiments alsoapply to peak-to-average power ratio (PAPR) reduction, and use of theterm PAR herein is intended to include at least PAR and PAPR. PAR is ametric that facilitates an assessment of the dynamic range of a signal,and a signal with a low PAR may be preferable, because it may allow thepower amplifier 316 to operate at higher power efficiencies withoutsubstantial signal distortion. In an embodiment, the PAR for each of thecandidate signals 342 may be calculated according to the followingequation:

$\begin{matrix}{{P\; A\; R\left\{ {y^{(d)}\lbrack n\rbrack} \right\}} = {\frac{\max_{n}{{y^{(d)}\lbrack n\rbrack}}^{2}}{E\left\lbrack {{y^{(d)}\lbrack n\rbrack}}^{2} \right\rbrack}.}} & \left( {{Equation}\mspace{14mu} 4} \right)\end{matrix}$

In an embodiment, signal selector 312 performs a next step of a selectedmapping (SLM) process, which is a PAR reduction tool that may reduce thePAR of OFDM symbols by multiple decibels (dBs). In a particularembodiment, signal selector 312 is adapted to identify the selectedsignal 346 as the candidate signal 342 with the lowest PAR. A selectedmapping (SLM) index, {tilde over (d)}, of the candidate signal 342 withthe lowest PAR may be determined, in an embodiment, according to thefollowing equation:

$\begin{matrix}{\overset{\sim}{d} = {\min\limits_{d}{P\; A\; R{\left\{ {y^{d}\lbrack n\rbrack} \right\}.}}}} & \left( {{Equation}\mspace{14mu} 5} \right)\end{matrix}$

In accordance with an embodiment, PAR reduction is achieved by using Dcandidate signals 342, and selecting the candidate signal 342 with thelowest PAR. In another embodiment, additional PAR reduction also may beachieved based on the design of the SPSs 330. More particularly, whenIDFT{S_(k) ^((d))}=s^((d))[n] has low PAR, the combined sequence ofy^((d))[n]=x^((d))[n]√{square root over (1−ρ)}+s^((d))[n]√{square rootover ((ρ))} may, on average, have a lower PAR than x^((d))[n]. Theextent of PAR reduction is related to the magnitude of the embeddingfactor, ρ. When the embedding factor is increased, PAR reductions alsoare increased. In an embodiment, the SPSs 330 are designed to have arelatively low PAR (e.g., PAR<0.5 dB). In a particular embodiment, theSPSs 330 are designed with arbitrary power spectral densities (PSD)using a convex optimization technique.

In order for the receiver (e.g., receiver 114, FIG. 1) to recover theinput data symbol 320, X_(k) (e.g., to determine an estimate,{circumflex over (X)}_(k), of the input data symbol), the receivershould have knowledge of or estimate the SLM index, {tilde over (d)}. Inan embodiment, the receiver has knowledge of possible values for S_(k)^((d)) and φ_(k) ^((d)) in the form of one or more tables that areaccessible to (e.g., stored at) the receiver (e.g., receiver 114), wherethose tables correspond to the phase shift value table and the SPS tableaccessible to the transmitter 300. Accordingly, when the receiver hasknowledge of SLM index, {tilde over (d)}, it may recover the input datasymbol 320, X_(k). Embodiments of methods and apparatus for a receiverto obtain knowledge of the SLM index, {tilde over (d)} (e.g., to recoverthe SLM index, {tilde over (d)}, or to determine an estimate {tilde over({circumflex over (d)} of the SLM index) will be discussed in moredetail below, in conjunction with FIG. 5. Basically, embodiments achieveblind phase sequence detection without time and/or frequencysynchronization, and/or a priori knowledge of the channel.

Up-converter 314 receives the selected signal 346, and is adapted toperform a frequency up-conversion and digital-to-analog conversionprocess on the selected signal 346 in order to convert the selectedsignal from a baseband or intermediate frequencies (IF) to the radiofrequency (RF) band. The analog up-converted signal 350 is thenamplified by power amplifier 316 to produce an amplified signal 352.Power amplifier 316 may add non-linear distortion to the amplifiedsignal 352. Accordingly, in an embodiment, transmitter 300 may include afeedback loop adapted to analyze the amplified signal 352 and to applydigital pre-distortion to the input data, although this is outside thescope of the present application and is not depicted in FIG. 3. Theamplified signal 352 is converted to an analog RF signal 360 andtransmitted over the channel (e.g., channel 106, FIG. 1) by antenna 318.Unlike some traditional techniques, the analog RF signal 360 may betransmitted without a preamble, and the embedded synchronization/pilotsequence information provides a way for a receiver robustly tosynchronize with a channel-affected version of the transmitted signal,as will be described in detail in conjunction with FIG. 5.

FIG. 4 is an example of a frequency-domain representation of a transmitsignal 400, in accordance with an example embodiment. Axis 402represents frequency, and axis 404 represents signal power (e.g., indB). Within frequency band 406, the transmit signal 400 includes a datacomponent 408 and an SPS component 410, which are modulated onto aplurality, N, of sub-carriers. More particularly, the subcarriersoccupied by the data component 408, X_(k), of the transmit signal 400,may be decomposed into several non-overlapping parts: 1) data-bearingsubcarriers 412, which may be denoted by a set of indices K_(d); pilotsubcarriers 414, which may be denoted by a set of indices K_(p); andnull subcarriers 416, which may be denoted by the set of indices K_(n).In an embodiment, X_(k∉K) _(d) =0, so that the data component 408 of thetransmit signal 400 only contains energy in data-bearing subcarriers412. Null subcarriers 416 may be constrained, in an embodiment, to zeroto limit the amount of spectral regrowth that may encroach onneighboring channels. Pilot signals 420 may be defined as part of theSPS (e.g., SPS 332, FIG. 3 and SPS 538, FIG. 5). The subcarriersoccupied by the SPS component 410 of the transmit signal 400, may bedecomposed into the same non-overlapping parts as the data component408, or more particularly: 1) synchronization subcarriers 412, K_(d);pilot subcarriers 414, K_(p); and null subcarriers 416, K_(n). Thesesignal segmentations may be summarized as Table 1, below:

TABLE 1 k ε K_(d) k ε K_(p) k ε K_(n) X_(k) ≠0 =0 =0 S_(k) ≠0 ≠0 =0Y_(k) ≠0 ≠0 =0

Although fifty-one total sub-carriers (e.g., N=51), thirty-eightdata-bearing subcarriers 412, five pilot subcarriers 414, and eight nullsub-carriers 416 are illustrated in FIG. 4, these numbers are used forexample purposes only, and more or fewer total sub-carriers,data-bearing subcarriers 412, pilot subcarriers 414, and/or nullsub-carriers 416 may be utilized, in other embodiments.

SPS component 410 includes synchronization sequence information 422conveyed within synchronization subcarriers 412 (e.g., data-bearingsubcarriers 412), and a plurality of pilot signals 420 conveyed withinpilot subcarriers 414, in an embodiment. Because at least some of thesynchronization subcarriers 412 occupied by the SPS component 410 arethe same as the data-bearing subcarriers 412 occupied by the datacomponent 408, the synchronization sequence information 422 (and thusthe SPS component 410) may be considered to be “embedded” within thedata component 408.

Pilot signals 420 (or pilot subcarriers 414) have constant power and areevenly spaced (e.g., a same number of data-bearing subcarriers 414 existbetween consecutive pilot subcarriers 414), in an embodiment. Inalternate embodiments, the positioning and spacing of pilot signals 420may be different from that illustrated in FIG. 4. In a particularembodiment, the pilot subcarrier 414 spacing is less than the number ofnull subcarriers (e.g., N/|K_(p)|>|K_(n)|), which may result in arelatively small channel estimation mean square error (MSE). The amountof power in pilot subcarriers 414 may be quantified according to theequation:

$\begin{matrix}{{\beta = \frac{\sum\limits_{k \in K_{p}}\; {S_{k}}^{2}}{\sum\limits_{k \in {K_{p}\bigcup K_{d}}}\; {S_{k}}^{2}}},} & \left( {{Equation}\mspace{14mu} 6} \right)\end{matrix}$

which is the ratio of pilot power to the total SPS power.

Referring also to FIG. 3, in the frequency domain, transmit signal 400may be represented according to the equation:

Y _(k) =X _(k)√{square root over (1−ρ)}+S _(k)√{square root over(ρ)},  (Equation 7)

where X_(k) represents and input data symbol 320, S_(k) represents anSPS 332, √{square root over (1−ρ)} represents a first scaling factor322, and √{square root over (ρ)} represents a second scaling factor 330.As mentioned previously, because the first and second scaling factors322, 330 have an inverse relationship, the value of the embeddingfactor, p, dictates how much relative signal power is allocated to thedata component 408, X_(k), of the transmit signal 400 as opposed to theSPS component 410, S_(k), of the transmit signal 400.

In an embodiment, the embedding factor, ρ, has a fixed value, andaccordingly the first scaling factor 322 and the second scaling factor330 also have fixed values. In another embodiment, the transmitter 300may adjust the value of the embedding factor dynamically. When theembedding factor is increased, the relative power of the SPS component410 with respect to the data component 408 also will increase. This maybe desirable, for example, when the channel is relatively harsh, andincreased PAR reductions are desired. However, a tradeoff to increasingthe embedding factor (and thus increasing PAR reductions) may be that,as a harsh channel improves (e.g., becomes less harsh), the receiver mayhave more than enough channel SNR to demodulate, although the receivedsignal SNR may be limited by the distortion induced by the poweramplifier 316. In an embodiment, the receiver may feed back informationback to the transmitter 300, which indicates the receiver demodulationperformance, and thus that the transmitter 300 may increase D and/or ρ.Such increases may enable transmitter 300 further to reduce PAR and tominimize the probability of distortion to the transmitted signal thatmay be induced by the non-linear power amplifier 316. Alternatively,when the embedding factor is decreased, the relative power of the SPScomponent 410 with respect to the data-bearing component 408 also willdecrease. Decreasing the embedding factor may be desirable, for example,when the power amplifier 316 is not inducing significant distortion ontothe transmitted signal, and when the demodulation performance of thereceiver (e.g., as indicated through feedback from the receiver) is notsignificantly limited by power amplifier induced distortions and/or bychannel multi-path induced distortion, provided that sufficientsynchronization performance may still be achieved. However, decreasingthe embedding factor may result in smaller PAR reductions. In stillanother embodiment, the value of the embedding factor may be set to 0,and/or data/scaling factor combiner 302 and SPS/scaling factor combiners306 may be disabled. In that case, transmit signal 400 will include onlya data component 408, as the power of any SPS component 410 effectivelywill have been reduced to zero. In such an embodiment, a preamble (notillustrated) may be transmitted along with the data in order tofacilitate synchronization with the signal at the receiver.

FIG. 5 is a simplified block diagram of a receiver 500, in accordancewith an example embodiment. Receiver 500 includes an antenna 502, adown-converter 504, a plurality of correlators 506, a peak detector 508,offset estimator/corrector 509, a channel estimator/corrector 516, anSPS removal element 518, scaling element 520, and a phase shift element522 operatively coupled together as illustrated in FIG. 5, in anembodiment. As will be described in detail below, receiver 500 includesa conjugate correlation receiver, which is adapted to perform a blindphase sequence detection method, in which the receiver 500 may excludethe traditional processes of performing time or frequencysynchronization, and in which the receiver 500 may not have a prioriknowledge of the channel characteristics.

Antenna 502 is adapted to receive a wireless RF signal 530 from thechannel, and to produce an analog RF signal 532. As discussed in detailabove, the wireless RF signal 530 represents a channel-affected versionof a selected signal that includes a data signal combined with an SPS.Down-converter 532 is adapted to perform an analog-to-digital conversionand a frequency down-conversion process on the analog RF signal 532, inorder to produce an IF or baseband received signal 534. Essentially, thereceived signal 534 represents a channel-affected version of a selectedsignal (e.g., selected signal 346, FIG. 3) that was transmitted by atransmitter (e.g., transmitter 300, FIG. 3) over a channel. The receivedsignal 534 may be represented by the following equation:

z ^(({tilde over (d)})) [n]=(f _(PA)(y ^(({tilde over (d)})) [n−n₀])*h[τ])e ^(−j2πε/N) +η[n],  (Equation 8)

where f_(PA)(.) represents the power amplifier input-to-outputcharacteristic, which may be assumed to be time-invariant, h[τ]represents a multi-path fading component of the channel,y^(({tilde over (d)}))[n−n₀] represents the transmitted signal,y^(({tilde over (d)}))[n], subjected to a TO component, e^(−j2πε/N)represents a CFO component, η[n] represents an additive noisecomponent, * is the convolution operator, and {tilde over (d)} is theSLM index. It is to be noted that any carrier phase shift presentbetween the transmitter and receiver is assumed to be included in thephase of the channel at the receiver.

As will be described in detail below, transmitter 500 is adapted todetermine estimated channel perturbations (e.g., multi-path fading, TO,CFO, and/or other signal perturbations) reflected within the receivedsignal 534, to apply corrections to the received signal 534 based on theestimated channel perturbations, and to produce an output data symbol580 based on the corrected received signal, where the output data symbol580 represents an estimate of the input data symbol (e.g., input datasymbol 320, FIG. 3) that was processed and transmitted by thetransmitter.

In an embodiment, estimated channel perturbations are determined by theplurality of correlators 506, the peak detector 508, the offsetestimator/corrector 509, and the channel estimator/corrector 516. Theplurality of correlators 506 includes computational apparatus adapted toreceive the received signal 534, to obtain a plurality of candidatesynchronization sequences 538, and to produce a plurality of conjugatecorrelation (CC) outputs 536, r^((d))[u]. More particularly, eachcorrelator 506 is adapted to correlate the received signal 534 with adifferent candidate synchronization sequence 538, s^((d))[n].

In an embodiment, the candidate synchronization sequences 538 includetime-domain versions of the same synchronization/pilot sequences (e.g.,SPSs 332, FIG. 3) as were combined by the transmitter (e.g., transmitter300, FIG. 3) with the phase shifted input data (e.g., phase shiftedinput data 328, FIG. 3). As mentioned previously, both the transmitter(e.g., transmitter 300) and the receiver 500 each may have knowledge ofthe candidate SPSs by each having access to substantively identicaltables of SPSs, although the transmitter's SPS table may include SPSsrepresented in the frequency domain, and the receiver's SPS table mayinclude the same SPSs represented in the time domain, in an embodiment.

The plurality of conjugate correlation outputs 536 may be represented bythe equation:

r ^((d)) [u]=CC{s ^((d)) [n],z ^(({tilde over (d)})) [n−u]},  (Equation9)

where the conjugate correlation between two length-N sequences may bedefined as:

$\begin{matrix}{{C\; C\left\{ {{a\lbrack n\rbrack},{b\lbrack n\rbrack}} \right\}} = {\left( {\sum\limits_{n = 0}^{{({N/2})} - 1}{{a^{*}\lbrack n\rbrack}{b\left\lbrack {n - u} \right\rbrack}}} \right) \cdot {\left( {\sum\limits_{n = {N/2}}^{N - 1}{{a^{*}\lbrack n\rbrack}{b\left\lbrack {n - u} \right\rbrack}}} \right)^{*}.}}} & \left( {{Equation}\mspace{14mu} 10} \right)\end{matrix}$

In an embodiment, the number of conjugate correlation outputs 536produced equals the candidate number quantity D, although the number ofconjugate correlation outputs 536 may be different, in otherembodiments.

In an embodiment, the received signal 534 may be divided into aplurality of subcode sequences in order to reduce the number ofoperations associated with performing the correlation process. In suchan embodiment, each conjugate correlation output 536 may be produced, bygenerating a sub-correlation for each subcode sequence, and summingtogether the sub-correlations to form a summed result having a singlecorrelation peak.

Correlators 506 provide the plurality of conjugate correlation outputs536 to peak detector 508. In an embodiment, correlators 506 may notprovide (or peak detector 508 may not evaluate) those of the pluralityof conjugate correlation outputs 536 that have correlation peaks below athreshold. Peak detector 508 includes computational apparatus adapted todetermine an estimate of the SLM index 540, {tilde over ({circumflexover (d)}, based on the conjugate correlation outputs 536. In anembodiment, the SLM index estimate 540 is determined according to theequation:

$\begin{matrix}{\hat{\overset{\sim}{d}} = {\arg \; {\max\limits_{d}{{{r^{(d)}\lbrack u\rbrack}}.}}}} & \left( {{Equation}\mspace{14mu} 11} \right)\end{matrix}$

Accordingly, the SLM index estimate 540 corresponds to the conjugatecorrelation output 536 that represents a highest correlation peak.Unlike traditional methods, embodiments include blind phase sequencedetection criterion (e.g., no side information representing the SLMindex is transmitted) in order to determine the SLM index estimate 540,and the SLM index estimate 540 is determined based on the conjugatecorrelations between the received signal 534 and the candidatesynchronization sequences 538. Correct detection of d may depend on themagnitude of the peaks of |r^((d))[u]| for d≠{tilde over (d)}, alsoreferred to herein as “spurious peaks.” When the spurious peaks all areless than the peak in |r^(({tilde over (d)}))[u]|, {tilde over (d)} maybe correctly detected (e.g., {tilde over ({circumflex over (d)}={tildeover (d)}). In an embodiment, the candidate SPSs 538 are designed sothat the spurious peaks are low. In a particular embodiment, thecandidate SPSs 538 are designed so that:

[max CC{s ^((d)) [n],s ^((d)) [n−u]}]<th _(self),  (Equation 12)

where th_(self) is a threshold that provides adequate systemperformance. Peak detector 508 provides the SLM index estimate 540,{tilde over ({circumflex over (d)}, to offset estimator/corrector 509(or more particularly to coarse offset estimator 510), along with the{tilde over ({circumflex over (d)} th conjugate correlation output 541(although this may be obtained from elsewhere, as well).

Offset estimator/corrector 509 includes a coarse offset estimator 510,an offset corrector 512, a time domain-to-frequency domain (TD-to-FD)transformer 514, a fine offset estimator 515, and a frequencydomain-to-time domain (FD-to-TD) transformer 517, in an embodiment.Coarse offset estimator 510 includes computational apparatus adapted todetermine a plurality of channel perturbations, including coarse timingoffset (TO) estimates 542 and coarse carrier frequency offset (CFO)estimates 544.

In an embodiment, coarse offset estimator 510 is adapted to determine acoarse timing offset estimate 542, {circumflex over (n)}₀, according tothe equation:

$\begin{matrix}{{\hat{n}}_{0} = {\arg \; {\max\limits_{u}{{{r^{(\hat{\overset{\sim}{d}})}\lbrack u\rbrack}}.}}}} & \left( {{Equation}\mspace{14mu} 13} \right)\end{matrix}$

Accordingly, the coarse timing offset estimate 542 is determined basedon the maximum of the {tilde over ({circumflex over (d)} th conjugatecorrelation output. Assuming that {tilde over ({circumflex over(d)}={tilde over (d)}, the coarse timing offset estimate should bedetermined (or “detected”) correctly as long as|r^(({tilde over (d)}))[n₀]>r^(({tilde over (d)}))[n] for n≠n₀.

In an embodiment, coarse offset estimator 510 also is adapted todetermine a coarse estimate of the carrier frequency offset (CFO) 544,{circumflex over (ε)}, according to the equation:

{circumflex over (ε)}=angle(r ^(({tilde over ({circumflex over (d)})) [n₀])  (Equation 14)

Essentially, the coarse CFO estimate is determined as the phase of theconjugate correlation output 536 that was determined by peak detector508 to have the highest correlation peak.

In an embodiment, the coarse offset and estimator 510 provides theestimated channel perturbations (e.g., coarse timing offset estimates542 and coarse CFO estimates 544) to offset corrector 512. Offsetcorrector 512 includes computational apparatus adapted to receive thereceived signal 534 and the estimated channel perturbations, and toeffectively compensate for those estimated channel perturbations in thereceived signal 534 by aligning the received signal 534 on a symbolboundary using the coarse timing offset estimate 542 and the coarse CFOestimate 544, which may include removing the cyclic extension from thereceived signal 534. In an embodiment, offset corrector 512 produces acoarsely-corrected signal 550.

Once the coarse timing and carrier frequency offsets are removed, thecoarsely-corrected signal 550 may be converted to the frequency domainby time domain-to-frequency domain (TD-to-FD) transformer 514, whichincludes computational apparatus adapted to perform a timedomain-to-frequency domain transformation on the corrected signal 550,in order to produce a frequency-domain, coarsely-corrected signal 553.The time domain-to-frequency domain conversion may include a Fouriertransform (FT) or, more particularly, a fast Fourier transform (FFT), invarious embodiments, although other types of time domain-to-frequencydomain conversions may be performed in other embodiments.

In an embodiment, fine offset estimation may then be performed usingfine offset estimator 515. In an embodiment, fine offset estimator 515determines a fine CFO estimate, which is applied to thecoarsely-corrected signal 550 by offset corrector 512. In an embodiment,fine offset estimator 515 determines a fine CFO estimate, {circumflexover (ε)}, using the pilot signals (e.g., pilot signals 420, FIG. 4)within the frequency-domain, coarsely-corrected signal 553. In anembodiment, this includes estimating the phase of each pilot signal(e.g., pilot signals 420), and determining the phase change in anyparticular pilot signal from OFDM symbol to OFDM symbol. Thus, the fineCFO estimate may be determined using the common sub-carrier phasedifference between OFDM symbols, which may then be averaged across allpilot sub-carriers to minimize estimation variance.

The frequency domain pilot part of the received signal for twoconsecutive sets of pilot symbols may be approximated as Y_(k1)^(p)=X_(k1) ^(p)H_(k1) ^(p)e^(−j2πε′) ¹ ^(/N) and Y_(k2) ^(p)=X_(ki2)^(p)H_(k2) ^(p)e^(−j2πε′) ² ^(/N), respectively. The phases φ_(ki1) andφ_(ki2), may be computed as ∠Y_(k1) ^(p) and ∠Y_(k2) ^(p) (where ∠represents the angle), respectively. Then, the fine CFO estimate, may bedetermined according to the equation:

$\begin{matrix}{{C\; F\; O} = {\frac{1}{2\; \pi \; T_{s}}{\sum\limits_{k = 0}^{{K_{pi} - 1}}\; {\left( {\varphi_{k\; 1} - \varphi_{k\; 2}} \right).}}}} & \left( {{Equation}\mspace{14mu} 15} \right)\end{matrix}$

Fine offset estimator 515 may provide the fine CFO estimate to offsetcorrector 512 via a feedback path (not illustrated). In addition, fineoffset estimator 515 provides a feedback version 545 of thefrequency-domain, coarsely-corrected signal to offset corrector 512 viafrequency domain-to-time domain (FD-to-TD) transformer 517, whichtransforms the feedback version 545 of the coarsely-corrected signalinto the time domain to produce a time-domain, fed back,coarsely-corrected signal 547. In an alternate embodiment, thecoarsely-corrected signal 550 is retained in memory, and is not fed backto offset corrector 512. Either way, offset corrector 512 applies thefine CFO estimate to the coarsely-corrected signal (either signal 550 or547) to re-produce the finely-corrected signal 551. In an alternateembodiment, fine CFO correction may be performed in the frequency domainafter fine offset estimator 515, rather than performing the fine CFOcorrection in the time domain by offset corrector 512.

In a further embodiment, fine offset estimator 515 also may determine afine timing offset estimate and/or a carrier phase offset estimate. Forexample, fine offset estimator 515 may determine a fine timing offsetestimate based on the phase slope between pilot sub-carriers common toeach OFDM symbol, which also can be averaged over all symbols. Fineoffset estimator 515 may determine a carrier phase offset estimate fromthe mean value of the phase slope in each OFDM symbol, in an embodiment.

When a fine timing and/or carrier phase offset are estimated, fineoffset estimator 515 provides the fine timing and/or carrier phaseoffsets to channel estimator/corrector 516, in an embodiment, forcorrection of the fine timing and/or carrier phase offset in thefrequency domain. In an alternate embodiment, fine offset estimator 515may provide the fine timing and/or carrier phase offsets, if estimated,to offset corrector 512 for correction in the time domain.

Either way, the finely-corrected signal 551 is transformed to thefrequency domain by TD-to-FD transformer 514, and the resultingcorrected signal 552 is provided to channel estimator/corrector 516.Channel estimator/corrector 516 receives the corrected signal 552,determines a channel estimate, and based on the channel estimate,proceeds to equalize the channel effects in the corrected signal 552 toproduce an equalized combined signal 554. Channel estimator/corrector516 is adapted to determine a channel estimate, Ĥ_(k) based on thecorrected signal 552. In an embodiment, the channel estimate isdetermined by generating a first quantity according to the equation:

W _(k) ^(({tilde over (d)})) =IDFT{z ^(({tilde over (d)}))[n+{circumflex over (n)} ₀ ]}e ^(j2π{circumflex over (ε)}/N),  (Equation16)

which yields W_(k) ^(({tilde over (d)}))=Y_(k)^(({tilde over (d)}))H_(k)+η_(k)+δ_(k)+t_(k), where δ_(k) is thedistortion noise caused by the power amplifier (e.g., power amplifier316, FIG. 3), t_(k) is the inter-carrier interference, and H_(k) andη_(k) are the IDFTs of h[n] and η[n], respectively. From W_(k)^(({tilde over (d)})), channel estimator/corrector 516 may estimate thechannel in the pilot subcarriers (e.g., pilot subcarriers 414, FIG. 4)according to the equation:

$\begin{matrix}{{{\hat{H}}_{k} = \frac{W_{k}^{(\overset{\sim}{d})}}{S_{k}^{(\hat{\overset{\sim}{d}})}\sqrt{\rho}}},{k \in {K_{p}.}}} & \left( {{Equation}\mspace{14mu} 17} \right)\end{matrix}$

In an embodiment, channel estimator/corrector 516 may interpolate thepilot subcarrier channel estimates to the data-bearing subcarriers(e.g., data-bearing subcarriers 412, FIG. 4), kεK_(d) so that Ĥ_(k) isdefined for kεK_(d)∪K_(p).

In an alternate embodiment, assumptions may be made that all of thesynchronization works perfectly (e.g., {tilde over ({circumflex over(d)}={tilde over (d)}, {circumflex over (n)}₀=n₀, and {circumflex over(ε)}=ε) and that no distortion noise is introduced by the transmitterpower amplifier (e.g., power amplifier 316, FIG. 3). With thoseassumptions, the first quantity represented in Equation 16, above, maybe simplified to:

W _(k) ^(({tilde over (d)})) =Y _(k) ^(({tilde over (d)})) H_(k)+η_(k),  (Equation 18)

where η_(k)≈CN(0,σ_(η) ²). Using these assumptions and the first orderapproximation that E[|η_(k)|²|{circumflex over (X)}_(k)|²H_(k)]≈σ² forkεK_(d), the symbol estimate MSE may be determined according to theequation:

$\begin{matrix}{{E\left\lbrack {{{{{\hat{X}}_{k} - X_{k}}^{2}}}H_{k}} \right\rbrack} \approx {\frac{\sigma^{2}}{{H_{k}}^{2}} \cdot {\begin{pmatrix}{\frac{\left( {1 - \beta} \right){K_{p}}}{{\beta \left( {1 - \rho} \right)}{K_{d}}} +} \\{\frac{K_{p}}{\beta \; \rho {K_{d}}} + \frac{1}{1 - \rho}}\end{pmatrix}.}}} & \left( {{Equation}\mspace{14mu} 19} \right)\end{matrix}$

As Equation 19 indicates, the MSE is dependent on the ratio of pilot todata subcarriers

$\frac{K_{p}}{K_{d}}.$

Also, the minimizing the pilot subcarrier power is achieved by settingβ=1 when perfect synchronization is assumed. However, in an embodiment,β is selected such that β<1, in order to achieve desired synchronizationperformance.

Channel estimator/corrector 516 may then generate an equalized combinedsignal 554 by equalizing the channel effects based on the channelestimate. After the various offset and channel corrections, theequalized combined signal 554 may be represented as:

z ^(({tilde over (d)})) [n]=((f _(PA)(y ^(({tilde over (d)})) [n−n₀])*h[τ])e ^(−j2πε/N+) η[n])e ^(j2π{circumflex over (ε)}/N)  (Equation20)

SPS removal element 518 includes computational apparatus adapted toreceive the equalized combined signal 554, and to remove the scaled SPS562 corresponding to the SLM index estimate 540 from the equalizedcombined signal 554 (e.g., to combine −√{square root over (ρ)}s_(k)^(({tilde over ({circumflex over (d)})) with the equalized combinedsignal 554) in order to produce an estimated, phase shifted data signal564. Scaling element 520 is adapted to apply a scaling factor to theestimated, phase shifted data signal 564, in order to produce a scaled,phase shifted data signal 566, which has a peak amplitude approximatelyequal to that of the original input data, X[n].

Phase shift element 522 includes computational apparatus adapted tophase shift the scaled, phase shifted data signal 566 by a phase shiftvalue 568 corresponding to the SLM index estimate 540 (e.g., to shiftthe scaled, phase shifted data signal 566 by e−jφ^(({tilde over ({circumflex over (d)})) ) The remaining signal isdemodulated in order to produce the output data symbol 580, {circumflexover (X)}_(k)[n]. When the SLM index estimate 540 represents acorrectly-detected SLM index (e.g., an SLM index corresponding to theselected signal 346, FIG. 3, identified at the transmitter 300), thenblind phase sequence detection has been robustly performed by receiver500, and the output data symbol 580 reflects an accurate estimate of theinput data symbol (e.g., input data symbol 320, FIG. 3).

FIG. 6 is a flowchart of a method for generating and transmittingwireless signals that include embedded synchronization/pilot sequences,in accordance with an example embodiment. Embodiments of the method areonly briefly discussed in conjunction with FIG. 6, as various detailsand alternate embodiments were discussed in more detail above.

Referring also to FIG. 3, the method may begin, in block 602, when atransmitter (e.g., transmitter 300) receives (e.g., by data/scalingfactor combiner 302) an input data symbol (e.g., input data symbol 320).In block 604, a first scaling factor (e.g., first scaling factor 322)may be applied to the input data symbol, in order to produce a scaledinput data symbol (e.g., scaled input data symbol 324). As discussedpreviously, the first scaling factor may have a value of √{square rootover (1−ρ)}, where ρ is an embedding factor having an absolute valuebetween 0 and 1. In other embodiments, the first scaling factor may havea different value. In block 606, various different phase shifts (e.g.,phase shifts 326) are applied (e.g., by phase shifters 304) to thescaled input data symbol, in order to produce a plurality of phaseshifted input data signals (e.g., phase shifted input data signals 328).

In block 608, a plurality of SPSs (e.g., SPSs 332) are obtained, and asecond scaling factor (e.g., second scaling factor 330) is applied tothe plurality of SPSs in order to produce a plurality of scaled SPSs(e.g., scaled SPSs 334). As discussed previously, the second scalingfactor may have a value of √{square root over (ρ)}, in an embodiment,although the second scaling factor may have a different value, in otherembodiments. Preferably, but not essentially, the second scaling factorhas an inverse mathematical relationship with the first scaling factor(e.g., by varying the value of the embedding factor, as the secondscaling factor value increases, the first scaling factor valuedecreases, and vice versa).

In block 610, each one of the plurality of phase shifted input datasignals is combined (e.g., by data/SPS combiners 308) with one of thescaled SPSs in order to produce a plurality of combined signals (e.g.,combined signals 340). In block 612, a frequency domain-to-time domainconversion is performed (e.g., by FD-to-TD transformers 310) on each ofthe combined signals, in order to produce a plurality of candidatesignals (e.g., candidate signals 342).

In block 614, peak-to-average ratios (PARs) are determined (e.g., bysignal selector 312) for some or all of the candidate signals, and basedon the peak-to-average ratios, a selected signal (e.g., selected signal346) is identified from the candidate signals. As discussed previously,the selected signal may be identified as the candidate signal with thelowest PAR, in an embodiment. In block 616, the selected signal isup-converted (e.g., by up-converter 314), amplified (e.g., by poweramplifier 316), and transmitted over the channel (e.g., channel 106,FIG. 1). Although not illustrated or discussed herein, those of skill inthe art would realize that various other processes for conditioning,filtering, and/or processing the various signals prior to transmissionalso may be performed at various stages within the process of generatingand transmitting the selected signal. Upon transmitting the selectedsignal, the method may then end.

FIG. 7 is a flowchart of a method for receiving and processing wirelesssignals that include embedded synchronization/pilot sequences, inaccordance with an example embodiment. Embodiments of the method areonly briefly discussed in conjunction with FIG. 7, as various detailsand alternate embodiments were discussed in more detail above.

Referring also to FIG. 5, the method may begin, in block 702, when areceiver (e.g., receiver 500) receives (e.g., via antenna 502) awireless RF signal (e.g., RF signal 530) from the channel. The receivedRF signal includes a channel-affected version of a data signal combinedwith an SPS, as discussed in conjunction with the description ofembodiments of the transmitter (e.g., transmitter 300, FIG. 3), andembodiments of the method for generating and transmitting the wirelessRF signal (e.g., FIG. 6). In block 704, the received RF signal isdown-converted and digitized (e.g., by down-converter 532), in order toproduce an IF or baseband received signal (e.g., received signal 534).

In block 706, the received signal is correlated (e.g., by correlators506) with a plurality of SPSs (e.g., SPSs 538) to produce a plurality ofconjugate correlation outputs (e.g., conjugate correlation outputs 536).In block 708, an SLM index estimate (e.g., SLM index estimate 540) isdetermined (e.g., by peak detector 508), based on the conjugatecorrelation outputs.

In block 710, coarse offset estimates (e.g., coarse TO and coarse CFO)may be determined (e.g., by coarse offset estimator 510) based on theconjugate correlation output corresponding to the SLM index estimate. Inblock 712, corrections are made (e.g., by offset corrector 512) for thecoarse timing and carrier frequency offsets in the received signal, inorder to produce a coarsely-corrected signal (e.g., coarsely-correctedsignal 550). In block 714, fine estimated offsets (e.g., fine CFO, fineTO, and/or phase offset) may be determined (e.g., by fine offsetestimator 515) based on the coarsely-corrected signal, and in block 716,additional corrections may be made (e.g., by offset corrector 512 in thetime domain or by a frequency-domain offset corrector), in order toproduce a finely-corrected signal (e.g., finely-corrected signal 551).

In block 718, channel effects are estimated (e.g., by channelestimator/corrector 516) from a frequency-domain version of thefinely-corrected signal. The finely-corrected signal is then equalizedbased on the estimated channel effects, in order to produce an equalizedcombined signal (e.g., equalized combined signal 554).

In block 720, a scaled SPS (e.g., scaled SPS 562) corresponding to theSLM index estimate is removed (e.g., by SPS removal element 518) fromthe equalized combined signal, in order to produce an estimated, phaseshifted data signal (e.g., estimated, phase shifted data signal 564),which may be scaled (e.g., by scaling element 520). A phase shiftoperation is performed (e.g., by phase shift element 522), in block 722,which includes phase shifting the scaled, phase shifted data signal by aphase shift value corresponding to the SLM index estimate. Thisoperation results in the production of an output data symbol (e.g.,output data symbol 580), which reflects and estimate of the input datasymbol (e.g., input data symbol 320, FIG. 3). The method may then end.

FIGS. 8 and 9 indicate potential simulated results for systems thatemploy various example embodiments. For example, FIG. 8 is a chart 800plotting bit error rate (BER) performance that may be achieved using twoexample SPS-SLM embodiments. Chart 800 includes a signal-to-noise ratio(SNR) axis 802 and a BER axis 804. The SPSs used to generate plot 800were generated using a convex optimization technique, according to anembodiment. In addition, an ideal soft limiter channel was used with aninput backoff of 3 dB. The CFO was set to a constant (ε=0.2, where ε isa function of the subcarrier spacing, 1/T_(s), and number ofsubcarriers, N, where T_(s)=NT, and T is the baseband sampling period:Thus, an ε=0.2 represents a carrier frequency offset of 20% of thesubcarrier spacing), the multi-path channel was set to length 16 with anexponential delay spread such that AΣ_(τ=0) ¹⁴e^(−τ). Also, N=256, with|K_(p)|=16, |K_(d)|=240, and |K_(n)|=0. The pilot tones were evenlyspaced with equal power, and the embedding factor was chosen to beρ=0.35.

Trace 810 plots BER performance for a system in which embeddedsynchronization was used without any PAR reduction considerations (e.g.,the candidate number quantity, D=1, and S_(k) ⁽¹⁾ was generated with aprescribed power profile but random phases). In contrast, trace 812plots BER performance for a system in which PAR reduction was achievedusing SPS designed in accordance with a particular embodiment, and witha candidate number quantity, D=1 (no selective mapping). A comparisonbetween plots 810 and 812 indicates that significantly improved BERperformance may be achieved when PAR reduction is used, in accordancewith an example embodiment. Trace 814 plots BER performance for a systemin which SPS-SLM was used with PAR reduction, in accordance with variousembodiments, and with a candidate number quantity, D=8. A furthercomparison between plots 812 and 814 indicates that even furtherimproved BER performance may be achieved when the candidate numberquantity, D, is increased.

FIG. 9 is a chart 900 comparing PAR characteristics for signals producedusing various embedding factors, ρ, when the candidate number quantity,D, is selected as D=1 (no selective mapping) and D=8, in accordance withvarious example embodiments. Traces 901, 902, 903, and 904 plot PARcharacteristics for D=1 and for embedding factors of ρ=0, ρ=0.3, ρ=0.5,and ρ=0.7, respectively. Traces 905, 906, 907, and 908 plot PARcharacteristics for D=8 and for embedding factors of ρ=0, ρ=0.3, ρ=0.5,and ρ=0.7, respectively. A comparison between the various groups oftraces, where group 910 corresponds to D=1 and group 912 corresponds toD=8, and also to the difference between corresponding traces having thesame embedding factor value (e.g., traces 904 and 908) indicate thatsignificant PAR reductions may be achieved using SPS-SLM, in accordancewith various embodiments.

FIGS. 8 and 9 illustrate that embodiments may provide one or moresignificant benefits over traditional methods and apparatus. Forexample, embodiments may provide improved bandwidth efficiency oversystems that transmit a preamble for use in synchronization. Inaddition, use of various embodiments may result in significantly reducedPAR using SPS-SLM. Significant BER improvements also may be possibleusing SPS-SLM embodiments in harsh channel environments.

Embodiments of methods and apparatus for wirelessly communicatingsignals that include combined (e.g., embedded) synchronization/pilotsequences have now been described above. The foregoing detaileddescription is merely exemplary in nature and is not intended to limitthe inventive subject matter or the application and uses of theinventive subject matter to the described embodiments. Furthermore,there is no intention to be bound by any theory presented in thepreceding background or detailed description.

Those of skill in the art will recognize, based on the descriptionherein, that various other apparatus and processes may be included inembodiments of the systems and methods described herein forconditioning, filtering, amplifying, and/or otherwise processing thevarious signals. In addition, the sequence of the text in any of theclaims does not imply that process steps must be performed in a temporalor logical order according to such sequence unless it is specificallydefined by the language of the claim. The process steps may beinterchanged in any order, and/or may be performed in parallel, withoutdeparting from the scope of the inventive subject matter. In addition,it is to be understood that information within the various differentmessages, which are described above as being exchanged between thesystem elements, may be combined together into single messages, and/orthe information within a particular message may be separated intomultiple messages. Further, messages may be sent by system elements insequences that are different from the sequences described above.Furthermore, words such as “connected” or “coupled to” used indescribing a relationship between different elements do not imply that adirect physical connection must be made between these elements. Forexample, two elements may be connected to each other physically,electronically, logically, or in any other manner, through one or moreadditional elements, without departing from the scope of the inventivesubject matter.

Those of skill in the art would understand that information and signalsmay be represented using any of a variety of different technologies andtechniques. For example, data, instructions, commands, information,signals, bits, symbols, and chips that may be referenced throughout theabove description may be represented by voltages, currents,electromagnetic waves, magnetic fields or particles, optical fields orparticles, or any combination thereof.

Those of skill would further appreciate that the various illustrativelogical blocks, modules, circuits, and algorithm steps described inconnection with the embodiments disclosed herein may be implemented aselectronic hardware, computer software, or combinations of both. Toclearly illustrate this interchangeability of hardware and software,various illustrative components, blocks, modules, circuits, and stepshave been described above generally in terms of their functionality.Whether such functionality is implemented as hardware or softwaredepends upon the particular application and design constraints imposedon the overall system. Skilled technicians may implement the describedfunctionality in varying ways for each particular application, but suchimplementation decisions should not be interpreted as causing adeparture from the scope of the inventive subject matter.

The various illustrative logical blocks and modules described inconnection with the embodiments disclosed herein may be implemented orperformed with various types of computational apparatus, including butnot limited to, a general purpose processor, a digital signal processor(DSP), an application specific integrated circuit (ASIC), a fieldprogrammable gate array (FPGA) or other programmable logic device,discrete gate or transistor logic, discrete hardware components, or anycombination thereof designed to perform the functions described herein.A general-purpose processor may be a microprocessor, but in thealternative, the processor may be any conventional processor,controller, microcontroller, or state machine. A processor may also beimplemented as a combination of computing devices, such as a combinationof a DSP and a microprocessor, a plurality of microprocessors, one ormore microprocessors in conjunction with a DSP core, or any other suchconfiguration.

The steps of a method or algorithm described in connection with theembodiments disclosed herein may be embodied directly in hardware, inone or more software modules executed by a processor, or in acombination of the two. A software module may reside in random accessmemory, flash memory, read only memory (ROM), erasable programmable ROM(EPROM), electrical EPROM, registers, hard disk, a removable disk, acompact disc ROM (CD-ROM), or any other form of storage medium known inthe art. An exemplary storage medium is coupled to the processor suchthat the processor can read information from, and write information to,the storage medium. In the alternative, the storage medium may beintegral to the processor. The processor and the storage medium mayreside in an ASIC. The ASIC may reside in a user terminal. In thealternative, the processor and the storage medium may reside as discretecomponents in a user terminal.

An embodiment of a method for wirelessly communicating a signal includesthe steps of combining a plurality of phase shifted input data signalswith a plurality of synchronization/pilot sequences to produce aplurality of combined signals, performing frequency domain-to-timedomain transformations of the combined signals to produce a plurality ofcandidate signals, determining peak-to-average ratios for at least someof the plurality of candidate signals, identifying a selected signalfrom the plurality of candidate signals based on the peak-to-averageratios, and transmitting the selected signal over a wirelesscommunication channel.

In a further embodiment, identifying the selected signal includes thestep of identifying the selected signal as a signal of the plurality ofcandidate signals that has a lowest peak-to-average ratio of thepeak-to-average ratios. A further embodiment includes the steps ofreceiving an input data symbol, applying a first scaling factor to theinput data symbol to produce a scaled input data symbol, phase shiftingthe scaled input data symbol to produce the plurality of phase shiftedinput data signals, and prior to combining, applying a second scalingfactor to the plurality of synchronization/pilot sequences. In a furtherembodiment, applying the first scaling factor includes applying ascaling factor having a value of √{square root over (1−ρ)}, and applyingthe second scaling factor includes applying a scaling factor having avalue of √{square root over (ρ)}, where ρ is an embedding factor havinga value between 0 and 1. In a further embodiment, ρ is an embeddingfactor having a value in a range between about 0.25 and about 0.35. Afurther embodiment includes the step of obtaining the plurality ofsynchronization/pilot sequences from a set of pre-generatedsynchronization/pilot sequences, where each synchronization/pilotsequence of the plurality of synchronization/pilot sequences isidentified by a selective mapping (SLM) index. A further embodimentincludes the steps of receiving a received signal from the wirelesscommunication channel, where the received signal represents achannel-affected version of the selected signal, performing blind phasesequence detection in order to determine an estimate of a SLM index thatidentifies a synchronization/pilot sequence embedded within the selectedsignal, determining estimated channel perturbations within the receivedsignal based on the received signal and the estimate of the SLM index,applying corrections to the received signal, based on the estimatedchannel perturbations, to produce a corrected received signal, andproducing an output data symbol from the corrected received signal. In afurther embodiment, transmitting the selected signal includes the stepof transmitting the selected signal without a preamble.

Another embodiment of a method includes the steps of receiving areceived signal from a wireless communication channel, where thereceived signal represents a channel-affected version of a selectedsignal that was transmitted by a transmitter, and where the selectedsignal represents a signal selected by the transmitter from a pluralityof candidate signals, and where the received signal includes acombination of a phase shifted input data signal with asynchronization/pilot sequence. The method also includes determining anestimate of a selective mapping (SLM) index, which identifies thesynchronization/pilot sequence from a plurality of synchronization/pilotsequences, applying corrections to the received signal based onestimated channel perturbations within the received signal, whichestimated channel perturbations are determined based on the estimate ofthe SLM index, to produce an equalized combined signal, and producing anoutput data symbol from the equalized combined signal.

In a further embodiment, determining the estimate of the SLM indexincludes performing blind phase sequence detection in order to determinethe estimate of the SLM index. In a further embodiment, determining theestimate of the SLM index includes correlating the received signal witha plurality of synchronization/pilot sequences to produce a plurality ofconjugate correlation outputs, and determining the estimate of the SLMindex as an SLM index corresponding to the conjugate correlation outputhaving a highest correlation peak. In a further embodiment, correlatingthe received signal includes dividing the received signal into aplurality of subcode sequences, generating a sub-correlation for eachsubcode sequence of the plurality of subcode sequences, and summingtogether the sub-correlation for each subcode sequence to form aconjugate correlation output as a summed result having a singlecorrelation peak. In a further embodiment, applying the correctionsincludes the steps of determining a coarse timing offset based on themaximum of the conjugate correlation output having the highestcorrelation peak, determining a coarse carrier frequency offset from aphase of the conjugate correlation output having the highest correlationpeak, producing a coarsely-corrected signal by aligning the receivedsignal on a symbol boundary using the coarse timing offset and thecoarse frequency offset, producing a finely-corrected signal from thecoarsely-corrected signal and a fine carrier frequency offset determinedfrom the coarsely-corrected signal, determining a channel estimate basedon a frequency-domain version of the finely-corrected signal, andequalizing channel effects in the frequency-domain version of thefinely-corrected signal based on the channel estimate to produce theequalized combined signal. In a further embodiment, producing thefinely-corrected signal includes the steps of performing a timedomain-to-frequency domain transformation of the coarsely-correctedsignal to produce a frequency domain signal, determining the finecarrier frequency offset using pilot signals within the frequency domainsignal, performing a frequency domain-to-time domain transformation ofthe fine carrier frequency offset to produce a time-domain version ofthe fine carrier frequency offset, producing a time-domain version ofthe finely-corrected signal from the coarsely-corrected signal and thetime-domain version of the fine carrier frequency offset, and performinga time domain-to-frequency domain transformation of the time-domainversion of the finely-corrected signal to produce the finely-correctedsignal. In a further embodiment, producing the finely-corrected signalincludes the steps of performing a time domain-to-frequency domaintransformation of the coarsely-corrected signal to produce a frequencydomain signal, determining the fine carrier frequency offset using pilotsignals within the frequency domain signal, and correcting for the finecarrier frequency offset in the frequency domain to produce thefinely-corrected signal. In a further embodiment, producing thefinely-corrected signal includes producing the finely-corrected signalfrom the coarsely-corrected signal, the fine carrier frequency offset,and a fine timing offset determined from the coarsely-corrected signal.In a further embodiment, applying the corrections includes determiningone or more estimated channel perturbations selected from a group thatincludes a timing offset, a carrier frequency offset, and multi-pathfading. In a further embodiment, producing the output data includesremoving the synchronization/pilot sequence from the equalized combinedsignal to produce an estimated, phase shifted data signal, scaling theestimated, phase shifted data signal to produce a scaled, phase shifteddata signal, and phase shifting and demodulating the scaled, phaseshifted data signal to produce the output data symbol.

An embodiment of a system includes a transmitter and a receiver. Thetransmitter is adapted to combine a plurality of phase shifted inputdata signals with a plurality of synchronization/pilot sequences toproduce a plurality of combined signals, to perform frequencydomain-to-time domain transformations of the combined signals to producea plurality of candidate signals, to determine peak-to-average ratiosfor at least some of the plurality of candidate signals, to identify aselected signal from the plurality of candidate signals based on thepeak-to-average ratios, and to transmit the selected signal over awireless communication channel. The receiver is adapted to receive areceived signal from the wireless communication channel, where thereceived signal represents a channel-affected version of the selectedsignal, to perform blind phase sequence detection in order to determinean estimate of a SLM index that identifies a synchronization/pilotsequence embedded within the selected signal, to determine estimatedchannel perturbations within the received signal based on the receivedsignal and the estimate of the SLM index, to apply corrections to thereceived signal, based on the estimated channel perturbations, toproduce an equalized combined signal, and to produce an output datasymbol from the equalized combined signal.

In a further embodiment, the transmitter and the receiver are includedwithin wireless communication devices selected from a group thatincludes a cellular telephone, a radio, an unmanned autonomous vehicle,a one-way pager, a two-way pager, a personal data assistant, a computer,a base station, a wireless transmitter, and a wireless transceiver.

An embodiment of a transmitter includes a plurality of combiners adaptedto combine a plurality of phase shifted input data signals with aplurality of synchronization/pilot sequences to produce a plurality ofcombined signals, a frequency domain-to-time domain transformer, adaptedto perform frequency domain-to-time domain transformations of thecombined signals to produce a plurality of candidate signals, a signalselector adapted to determine peak-to-average ratios for at least someof the plurality of candidate signals, and to identify a selected signalfrom the plurality of candidate signals based on the peak-to-averageratios, and an antenna adapted to transmit the selected signal over awireless communication channel.

A further embodiment includes a first combiner adapted to apply a firstscaling factor to the input data symbol to produce a scaled input datasymbol, and a plurality of second combiners adapted to apply a secondscaling factor to the plurality of synchronization/pilot sequences,where the first scaling factor and the second scaling factor have aninverse relationship. A further embodiment includes a plurality of phaseshifters adapted to phase shift the input data symbol to produce theplurality of phase shifted input data signals.

An embodiment of a receiver includes a plurality of correlators, a peakdetector, an offset estimator/corrector, a channel estimator/corrector,a combiner, a scaling element, and a phase shifter. The plurality ofcorrelators is adapted to receive a received signal from a wirelesscommunication channel, where the received signal represents achannel-affected version of a selected signal that was transmitted by atransmitter, and where the selected signal represents a signal selectedby the transmitter from a plurality of candidate signals, and where thereceived signal includes a combination of a phase shifted input datasignal with a synchronization/pilot sequence. The peak detector isadapted to determine an estimate of a selective mapping (SLM) index,which identifies the synchronization/pilot sequence from a plurality ofsynchronization/pilot sequences. The offset estimator/corrector isadapted to determine, based on the received signal and the estimate ofthe SLM index, a coarse timing offset estimate, a coarse carrierfrequency offset estimate, and a fine carrier frequency offset estimate,and to correct the received signal using the coarse timing offsetestimate, a coarse carrier frequency offset estimate, and a fine carrierfrequency offset estimate to produce a finely-corrected signal. Thechannel estimator/corrector is adapted to determine a channel estimatebased on a frequency-domain version of the finely-corrected signal, andto equalize channel effects in the frequency-domain version of thefinely-corrected signal based on the channel estimate to produce anequalized combined signal. The combiner is adapted to remove thesynchronization/pilot sequence from the equalized combined signal toproduce an estimated, phase shifted data signal. The scaling element isadapted to scale the estimated, phase shifted data signal to produce ascaled, phase shifted data signal, and the phase shifter and demodulatorare adapted to phase shift and demodulate the scaled, phase shifted datasignal to produce the output data symbol.

While various exemplary embodiments have been presented in the foregoingdetailed description, it should be appreciated that a vast number ofvariations exist. It should also be appreciated that the exemplaryembodiments are only examples, and are not intended to limit the scope,applicability or configuration of the inventive subject matter in anyway. Rather, the foregoing detailed description will provide thoseskilled in the art with a convenient road map for implementing variousembodiments of the inventive subject matter, it being understood thatvarious changes may be made in the function and arrangement of elementsdescribed in an exemplary embodiment without departing from the scope ofthe inventive subject matter as set forth in the appended claims andtheir legal equivalents.

1. A method for wirelessly communicating a signal, the method comprising the steps of: combining a plurality of phase shifted input data signals with a plurality of synchronization/pilot sequences to produce a plurality of combined signals; performing frequency domain-to-time domain transformations of the combined signals to produce a plurality of candidate signals; determining peak-to-average ratios for at least some of the plurality of candidate signals; identifying a selected signal from the plurality of candidate signals based on the peak-to-average ratios; and transmitting the selected signal over a wireless communication channel.
 2. The method of claim 1, wherein identifying the selected signal comprises the step of: identifying the selected signal as a signal of the plurality of candidate signals that has a lowest peak-to-average ratio of the peak-to-average ratios.
 3. The method of claim 1, further comprising: receiving an input data symbol; applying a first scaling factor to the input data symbol to produce a scaled input data symbol; phase shifting the scaled input data symbol to produce the plurality of phase shifted input data signals; and prior to combining, applying a second scaling factor to the plurality of synchronization/pilot sequences.
 4. The method of claim 3, wherein applying the first scaling factor comprises applying a scaling factor having a value of √{square root over (1−ρ)}, and wherein applying the second scaling factor comprises applying a scaling factor having a value of √{square root over (ρ)}, wherein ρ is an embedding factor having a value between 0 and
 1. 5. The method of claim 4, wherein ρ is an embedding factor having a value in a range between about 0.25 and about 0.35.
 6. The method of claim 1, further comprising: obtaining the plurality of synchronization/pilot sequences from a set of pre-generated synchronization/pilot sequences, wherein each synchronization/pilot sequence of the plurality of synchronization/pilot sequences is identified by a selective mapping (SLM) index.
 7. The method of claim 1, further comprising the steps of: receiving a received signal from the wireless communication channel, wherein the received signal represents a channel-affected version of the selected signal; performing blind phase sequence detection in order to determine an estimate of a SLM index that identifies a synchronization/pilot sequence embedded within the selected signal; determining estimated channel perturbations within the received signal based on the received signal and the estimate of the SLM index; applying corrections to the received signal, based on the estimated channel perturbations, to produce a corrected received signal; and producing an output data symbol from the corrected received signal.
 8. The method of claim 1, wherein transmitting the selected signal comprises the step of: transmitting the selected signal without a preamble.
 9. A method comprising the steps of: receiving a received signal from a wireless communication channel, wherein the received signal represents a channel-affected version of a selected signal that was transmitted by a transmitter, and wherein the selected signal represents a signal selected by the transmitter from a plurality of candidate signals, and wherein the received signal includes a combination of a phase shifted input data signal with a synchronization/pilot sequence; determining an estimate of a selective mapping (SLM) index, which identifies the synchronization/pilot sequence from a plurality of synchronization/pilot sequences; applying corrections to the received signal based on estimated channel perturbations within the received signal, which estimated channel perturbations are determined based on the estimate of the SLM index, to produce an equalized combined signal; and producing an output data symbol from the equalized combined signal.
 10. The method of claim 10, wherein determining the estimate of the SLM index comprises: performing blind phase sequence detection in order to determine the estimate of the SLM index.
 11. The method of claim 10, wherein determining the estimate of the SLM index comprises: correlating the received signal with a plurality of synchronization/pilot sequences to produce a plurality of conjugate correlation outputs; and determining the estimate of the SLM index as an SLM index corresponding to the conjugate correlation output having a highest correlation peak.
 12. The method of claim 11, wherein correlating the received signal comprises: dividing the received signal into a plurality of subcode sequences; generating a sub-correlation for each subcode sequence of the plurality of subcode sequences; and summing together the sub-correlation for each subcode sequence to form a conjugate correlation output as a summed result having a single correlation peak.
 13. The method of claim 11, wherein applying the corrections comprises the steps of: determining a coarse timing offset based on the maximum of the conjugate correlation output having the highest correlation peak; determining a coarse carrier frequency offset from a phase of the conjugate correlation output having the highest correlation peak; producing a coarsely-corrected signal by aligning the received signal on a symbol boundary using the coarse timing offset and the coarse frequency offset; producing a finely-corrected signal from the coarsely-corrected signal and a fine carrier frequency offset determined from the coarsely-corrected signal; determining a channel estimate based on a frequency-domain version of the finely-corrected signal; and equalizing channel effects in the frequency-domain version of the finely-corrected signal based on the channel estimate to produce the equalized combined signal.
 14. The method of claim 13, wherein producing the finely-corrected signal comprises the steps of: performing a time domain-to-frequency domain transformation of the coarsely-corrected signal to produce a frequency domain signal; determining the fine carrier frequency offset using pilot signals within the frequency domain signal; performing a frequency domain-to-time domain transformation of the fine carrier frequency offset to produce a time-domain version of the fine carrier frequency offset; producing a time-domain version of the finely-corrected signal from the coarsely-corrected signal and the time-domain version of the fine carrier frequency offset; and performing a time domain-to-frequency domain transformation of the time-domain version of the finely-corrected signal to produce the finely-corrected signal.
 15. The method of claim 13, wherein producing the finely-corrected signal comprises the steps of: performing a time domain-to-frequency domain transformation of the coarsely-corrected signal to produce a frequency domain signal; determining the fine carrier frequency offset using pilot signals within the frequency domain signal; and correcting for the fine carrier frequency offset in the frequency domain to produce the finely-corrected signal.
 16. The method of claim 13, wherein producing the finely-corrected signal comprises: producing the finely-corrected signal from the coarsely-corrected signal, the fine carrier frequency offset, and a fine timing offset determined from the coarsely-corrected signal.
 17. The method of claim 10, wherein applying the corrections comprises determining one or more estimated channel perturbations selected from a group that includes a timing offset, a carrier frequency offset, and multi-path fading.
 18. The method of claim 10, wherein producing the output data comprises: removing the synchronization/pilot sequence from the equalized combined signal to produce an estimated, phase shifted data signal; scaling the estimated, phase shifted data signal to produce a scaled, phase shifted data signal; and phase shifting and demodulating the scaled, phase shifted data signal to produce the output data symbol.
 19. A system comprising: a transmitter adapted to combine a plurality of phase shifted input data signals with a plurality of synchronization/pilot sequences to produce a plurality of combined signals, to perform frequency domain-to-time domain transformations of the combined signals to produce a plurality of candidate signals, to determine peak-to-average ratios for at least some of the plurality of candidate signals, to identify a selected signal from the plurality of candidate signals based on the peak-to-average ratios, and to transmit the selected signal over a wireless communication channel; and a receiver adapted to receive a received signal from the wireless communication channel, wherein the received signal represents a channel-affected version of the selected signal, to perform blind phase sequence detection in order to determine an estimate of a SLM index that identifies a synchronization/pilot sequence embedded within the selected signal, to determine estimated channel perturbations within the received signal based on the received signal and the estimate of the SLM index, to apply corrections to the received signal, based on the estimated channel perturbations, to produce an equalized combined signal, and to produce an output data symbol from the equalized combined signal.
 20. The system of claim 19, wherein the transmitter and the receiver are included within wireless communication devices selected from a group that includes a cellular telephone, a radio, an unmanned autonomous vehicle, a one-way pager, a two-way pager, a personal data assistant, a computer, a base station, a wireless transmitter, and a wireless transceiver.
 21. A transmitter comprising: a plurality of combiners adapted to combine a plurality of phase shifted input data signals with a plurality of synchronization/pilot sequences to produce a plurality of combined signals; a frequency domain-to-time domain transformer, adapted to perform frequency domain-to-time domain transformations of the combined signals to produce a plurality of candidate signals; a signal selector adapted to determine peak-to-average ratios for at least some of the plurality of candidate signals, and to identify a selected signal from the plurality of candidate signals based on the peak-to-average ratios; and an antenna adapted to transmit the selected signal over a wireless communication channel.
 22. The transmitter of claim 21, further comprising: a first combiner adapted to apply a first scaling factor to the input data symbol to produce a scaled input data symbol; and a plurality of second combiners adapted to apply a second scaling factor to the plurality of synchronization/pilot sequences, wherein the first scaling factor and the second scaling factor have an inverse relationship.
 23. The transmitter of claim 21, further comprising: a plurality of phase shifters adapted to phase shift the input data symbol to produce the plurality of phase shifted input data signals.
 24. A receiver comprising: a plurality of correlators adapted to receive a received signal from a wireless communication channel, wherein the received signal represents a channel-affected version of a selected signal that was transmitted by a transmitter, and wherein the selected signal represents a signal selected by the transmitter from a plurality of candidate signals, and wherein the received signal includes a combination of a phase shifted input data signal with a synchronization/pilot sequence; a peak detector adapted to determine an estimate of a selective mapping (SLM) index, which identifies the synchronization/pilot sequence from a plurality of synchronization/pilot sequences; an offset estimator/corrector adapted to determine, based on the received signal and the estimate of the SLM index, a coarse timing offset estimate, a coarse carrier frequency offset estimate, and a fine carrier frequency offset estimate, and to correct the received signal using the coarse timing offset estimate, a coarse carrier frequency offset estimate, and a fine carrier frequency offset estimate to produce a finely-corrected signal; a channel estimator/corrector adapted to determine a channel estimate based on a frequency-domain version of the finely-corrected signal, and to equalize channel effects in the frequency-domain version of the finely-corrected signal based on the channel estimate to produce an equalized combined signal; a combiner adapted to remove the synchronization/pilot sequence from the equalized combined signal to produce an estimated, phase shifted data signal; a scaling element adapted to scale the estimated, phase shifted data signal to produce a scaled, phase shifted data signal; and a phase shifter and demodulator adapted to phase shift and demodulate the scaled, phase shifted data signal to produce the output data symbol. 